Flyback eht and sawtooth current generator

ABSTRACT

A flyback EHT- and sawtoothcurrent-generator using a transformer and switching means connected thereto which conduct during a scan period and do not conduct during a flyback period, and partly parasitic reactances connected to the transformer. During the flyback period the generator has at least two resonant frequencies so chosen that a substantially oscillation-free scan is produced and during the scan it has at least one resonant frequency which is approximately equal to an integral multiple of the reciprocal value of the duration of the scan period.

United States Patent Van Gils Feb. 19, 1974 [54] FLYBACK EHT ANDSAWTOOTH 2,854,592 9/1958 Ruth 315/29 CURRENT R OR 3,546,629 12/1970Walker, Jr. 315/29 X 3,546,630 12/1970 Jordan 315/29 X [75] Inventor:Cornelis Johannes Maria Van Gils,

Emmasingel, Eindhoven, Netherlands [73] Assignee: U.S. PhilipsCorporation, New

York, NY.

[22] Filed: Nov. 29, 1972 [21] Appl. No.: 310,346

[30] Foreign Application Priority Data Dec. 17, 1971 Netherlands 7117322[52] US. Cl. 315/29 [51] Int. Cl. H01j 29/70 [58] Field of Search315/29, 28, 27 TD [56] References Cited UNITED STATES PATENTS 3,500,1163/1970 Rietveld et al. 315/27 TD Primary Examiner-Carl D. QuarforthAssistant ExaminerP. A. Nelson Attorney, Agent, or Firm-Frank R. Trifari[57] ABSTRACT A flyback EHT- and sawtoothcurrent-generator using atransformer and switching means connected thereto which conduct during ascan period and do not conduct during a flyback period, and partlyparasitic reactances connected to the transformer. During the flybackperiod the generator has at least two resonant frequencies so chosenthat a substantially oscillationfree scan is produced and during thescan it has at least one resonant frequency which is approximately equalto an integral multiple of the reciprocal value of the duration of thescan period.

7 Claims, 5 Drawing Figures PATENTEU FEB I 9 i SHEET 1 [IF 3 Fig.2

PAH-1min FEB i 91974 SHEET 3 or 3 FLYBACK EHT AND SAWTOOTH CURRENTGENERATOR The invention relates to a flyback EHT- and sawtooth currentgenerator particularly for television display apparatus, includingswitching means which are periodically non-conducting during a flybackperiod 1' are conducting during a scan period T 1- and a network havinginput terminals connected to the switching means, said networkcomprising a transformer having at least one primary winding andpossibly one or more coils connected thereto through which said sawtoothcurrent flows during the scan period, and a secondary winding to which arectifier circuit is connected for generating said EHT from the voltagepulses occurring during the flyback period at the secondary winding,said network, due to the leakage inductance present between thetransformer windings having a first resonant frequency fa during theflyback period which frequency is substantially equal to the expression:

in which K l and S is a correction factor which is equal to the relativereduction of the slope of the sawtooth current at the end of the scanperiod relative to this slope at the centre of the scan period, and asecond resonant frequency f which is substantially equal to the saidexpression for K is equal to an odd integer of more than 1.

It is known from the Netherlands Patent Specification 88020 and from thebook Televisie by P.Kerkhof and W.Werner, third edition, chapter Xlll touse a transformer having one or more primary windings and a secondarywinding in flyback EHT' and sawtooth-current-generators for televisiondisplay apparatus and to proportion the impedances of the circuitelements present, such as the transformer windings, the leakageinductance between the primary windings and the secondary winding, theinductances of coils which are generally connected to the primarywinding as well as the parasitic and non-parasitic capacitances, in sucha manner that hardly any or no free oscillations occur in the secondaryvoltage during the scan period. Such free oscillations have thedrawbacks that useful energy is lost, that this useless energy is mainlydissipated in the transformer so that overheating of the transformer mayoccur and that the switching means conducting during the scan period maybecome prematurely nonconducting in case of large free oscillations.

The above-mentioned literature shows that said circuit constitutes a4"'-order network during the flyback period with two resonantfrequencies and that the said free oscillations may be maintained low byproportioning the circuit elements in such a manner that these tworesonant frequencies f and f,, have very defined values which aredependent on the duration of the flyback period 1 and the duration ofthe scan period T 7. As is apparent from the expression given in thepreamble, the optimum value of f and f is also slightly dependent on theextent to which the slope of the sawtooth current flowing through thedeflection coils varies when the so-called S-correction of this currentis used.

Theoretically a scan period completely without oscillations can beobtained by exact progortioning of f ences, this cannot be achieved inpractice. These interfering influences are, inter alia:

l. The presence of parasitic reactances in the network which are sosmall on the one hand that they can 5 no longer be controlled but are onthe other hand sufficiently large to make a scan without oscillationsimpossible.

2. The presence of losses in the network during the flyback period. Anessential part thereof is constituted by useful energy which is derivedthrough the rectifier.

3. Tolerances in components and production spread which make itimpossible to render the frequencies faand f sufficiently accuratelyequal to values which are necessary for a scan without oscillations.

An object of the invention is to provide a proportioning measure withwhich in spite of the said interfering influences the occurring scanoscillations can be maintained very small and to this end the flybackEHT and sawtooth current generator according to the invention gralmultiple of the reciprocal value of the duration of the scan period.

The said network has a certain resonant frequency 7 f9 located betweenthe two parallel resonant frequencies fa and f,, at which the impedanceof the network at the input terminals is at a minimum (equal to zero incases of a network which is completely without losses). This is actuallythe resonant frequency of the generator during the scan period, hencewith conducting switching means. It is known that for an exact tuning ofthe flyback EHT generator to a scan which is without oscillations onlythe parallel resonant frequencies f and f,, are important and that theseries resonant frequency f is irrelevant. The location off has onlyinfluence on the shape of the flyback pulse occurring across the inputterminals of the network.

The invention is based on the recognition of the fact that the value ofthe frequencyf has also a very great influence on the amplitude of thescan oscillations still remaining as a result of the said interferinginfluences and that this amplitude is at a minimum if the value of f issubstantially equal to an integral multiple of the reciprocal value ofthe duration of the scan period.

A generator of this kind is preferably designed in such a manner thatthe quality of the generator during thescan period is more than 25 atthe said frequency fa- The Applicant's pending Patent application No.245,144, filed Apr. 18, 1972 describes flyback EHT- andsawtoothcurrent-generators in which the flyback equal to the givenexpression:

K/2 1-){1+ 4/11 K r/T- r) (l 8)} in which K is an odd positive integer,for example, 1, p y qgl 5, 7 respectively. In such a case the networkhas a first series resonant frequency f5, between fa and f at which theinput impedance of the network is at a minimum and a second seriesresonant frequency f between )2 and f at which one of the frequency fand one of the frequency fg It is found that independent of each otherfor a minimum amplitude of the fig, component the frequency f3, is to bechosen optimum, that is to say, approximately equal to an integralmultiple of the reciprocal value of the duration of the scanperio dandfor a minimum amplitude of the f,;, component the frequency f must beapproximately equal to an integral multia ple of the reciprocal value ofthe duration of the scan period.

On the other hand it is, however, found that the component of the lowestfrequency generally has a considerably larger amplitude than thecomponent of the higher frequency. For a flyback El-IT and sawtoothcurrent generator in which the said network has a third resonantfrequency f between f and f during the flyback period which resonantfrequency is at least substantially equal to the given expression for Kbeing equal to an odd integer, a considerable reduction of the scanoscillations thus already achieved if the frequency fig between f,, andf at which the impedance of the network at the input terminals is at aminimum is substantially equal to an integral multiple of the reciprocalvalue of the duration of the scan period.

As noted hereinbefore an optimum proportioning is achieved when the saidfrequency f,;, between f.

and f, at which the impedance of the network at the input terminals isat a minimum is also substantially equal to an integral multiple of thereciprocal value of the duration of the scan period.

The invention will be further described with reference to the Figuresshown in the drawings. In these drawings:

FIG. 1 shows a first embodiment of a flyback EI-IT- andsawtoothcurrent-generator for which the invention can be used.

FIG. 2 shows the equivalent circuit diagram of the embodiment of FIG. 1.

FIG. 3 shows a diagram to explain the present invention.

FIG. 4 shows a second embodiment of a flyback EHT- andsawtoothcurrent-generator for which the invention can be used and FIG. 5shows the equivalent circuit diagram of the embodiment of FIG. 4.

The embodiment of FIG. 1 shows a transformer 1 having a primary winding2, one or more auxiliary windings 3 rigidly coupled to the primarywinding, and a secondary winding 4. A tap 6 on the primary winding Atransistor 12 operating as a switch is provided between the upper end ofthe primary winding and ground and a capacitor 13 is connected inparallel with this transistor. Said secondary winding 4 is connected toground at one end and at the other end to a rectifier circuit consistingof a rectifier l4 and a smoothing capacitor 15; the EHT generated by therectifier is applied to the acceleration anode of a television displaytube not further shown.

Switching pulses which periodically cut off the transistor 12 at the endof each scan period are applied between the base and emitter oftransistor 12 through a separating transformer 18, a series inductancel9 and a parallel diode 20. The transistor 12 is a so-called slowswitching transistor and the elements 19 and 20 are included so as toaccelerate the switching off of the transistor at the end of the scanperiod.

FIG. 2 shows the simplified equivalent circuit diagram of the circuit ofFIG. 1. In this diagram E denotes 0 the voltage source 7, SW denotes theswitch constiis connected to the positive terminal ofa voltage supplytuted by transistor 12 and diode 20. C is the capacitance of thecapacitor 13 increased by the collector emitter capacitance of thetransistor and the transformed parasitic capacitances of the primarywinding, the auxiliary windings, the deflection coils and the linearitycorrector. L is the inductance of the primary winding and the deflectioncoils and the linearity corrector connected thereto, all transformed tothe terminals to which the switch is connected. L is the leakageinductance between secondary and primary windings and C is the parasiticcapacitance, of the secondary winding and the input capacitance of therectifier circuit likewise transformed to the terminals to which theswitch is connected.

During the scan period switch SW is closed. The voltage E from thevoltage supply source 7 is therefore present across capacitor C1 andalso across inductor Ll. As a result, a (sawtooth) current linearlyvarying with time will flow through the inductor Ll. When, as a resultof a pulse applied to the base electrode of transistor 12, switch SW isrendered non-conducting, free oscillations will occur in the network asa result of the magnetic energy present in L1. These oscillationsproduce pulsatory voltages V1 and V2, the so-called flyback pulsesacross capacitors Cl and C2, respectively. As soon as the flyback pulseacross Cl decreases to the value of the supply voltage -E, that is tosay, as soon as the collector potential of transistor 12 becomesnegative relative to ground, the collector-base pn-junction of thetransistor is in the forward direction and the next scan periodcommences. Switch SW in the equivalent circuit diagram of FIG. 2therefore closes automatically as soon as the flyback voltage presentacross this switch becomes equal to zero.

It is to be noted that the sawtooth current in the circuit diagram ofFIG. 1 flows during the first part of the scan period through the diode20, the base-collector junction of the transistor and subsequentlythrough the transformer and the deflection coils to the'voltage supplysource and thus feeds back energy to the voltage supply source. Sometime after the commencement of the scan period the base-emitter junctionof the transistor is rendered conducting by means of the pulses appliedto the base electrode of the transistor so that during the second partof the flyback period the sawtooth current now reversed in polarity canflow from the voltage supply source through the transformer and thedeflection coils and subsequently through the collector electrode andthe emitter electrode of the transistor to ground; then the voltagesupply source supplies energy to the network.

It is to be ensured that during the scan period only a sawtooth currentflows through L1 and that free oscillations do not occur as a result ofelectrical or magnetical energy present in the inductor L2 and capacitorC2. Such a scan period without oscillations is obtained if it is ensuredthat the current flowing through L2 is equal to zero during the entirescan period and therefore also at the commencement and the end of theflyback period and that the voltages across C2 is equal to the batteryvoltage E.

To satisfy this condition the following two relations are to apply foreach resonant frequency a of the flyback period network, thus of thenetwork in case of an open switch SW.

ba t/V a L1 and hence through the deflection coils, there applies as anapproximation:

1r(i' li (2 1-)/(T r) in which T 1- is the duration of the scan period.When, however, the deflection current has a slightly S-shaped characteras a result of the S-correction capacitor of FIG. 1 which is left out ofconsideration in the equivalent circuit diagram of FIG. 2, whichcharacterisgpnventional in television display apparatus, there appliesas an approximation:

in which S is the relative reduction of the slope of the deflectioncurrent at the end of the scan period relative to this slope in thecentre of the scan period. The above-mentioned condition for at thenchanges to:

with f a 01/2 fa (K/Z 1') art here follows: {1+ 4/(K 1r /T T) u a s)}on) as already noted hereinbefore the equations (I) and (II) and hencealso equation (III) must apply to each resonant frequency of the flybacknetwork in order to obtain a scan without oscillations.

The network of FIG. 2 has two resonant frequencies a 2 1r f and y 2 rr fduring the flyback period namely:

and to obtain a scan without oscillations the network is proportioned insuch a manner that f satisfies equation (III) for K l and also f,,satisfied equation (III), for example, K 3 or K 5, etc.

In addition to the frequenciesf and f the network has a thirdcharacteristic frequency f i.e. the frequency located between f and f atwhich the input impedance of the network is at a minimum (zero). For thenetwork of FIG. 2 this is the series resonant frequency of L2 and C2,thus B =(21rf,9) =(1/L2C2). At a given period of T and at a previouslychosen flyback period of'r the values off and f are laid down by thecondition given in equation (III). The value f may however, be freelychosen between f, and 2.

It is known that for values l and f determined in accordance withequation (III) the value of ffi determines the shape of the flybackpulse occurring across the input terminals of the network. This isfurther shown in FIG. 3 by the curves denoted by V1 which are found fordifferent values off As this Figure shows, the voltage V] decreases forlow values of f before the termination of the flyback period to belowthe value of the battery voltage. In order to prevent this,f isgenerally chosen to be sufficiently large and particularlyf is locatedin the region Vf f, ffl It will be evident that it is chosen to be inthe region f f f f,, switching means are to be used in the flyback EHTand sawtooth current generator in such a manner that they do not conductbefore the termination of the previously determined flyback period 1-.

It has been found that the location of the frequency f has also aconsiderable influence on the amplitude of the oscillations occurringduring the scan period which oscillations are produced because a scanperiod which is completely without oscillations cannot be achieved dueto the interfering influences described in the preamble. This is furthershown in FIG. 3 by the curve ER. This curve shows as a function off theratio between the energy which gives rise to t e said unwantedoscillation during the scan period and the total electromagnetic energyduring the flyback period in the network. It is clearly shown that thisenergy ratio at given values off has maximum values and has inimumvalues at values located therebetween.

It has been found that the maximum values in the ER curve occur atvalues of f for which there applies rrf (T- 'r) tangent (rrf (T 1))However, since vrf (T 1-) is large relative to l it may be assumed thatsaid maximum values occur approximately at This is the case iff (T r)n+%(n integer). The minimum values located between the maximum valuesoccur at those values off for which there applies that f (T 'r) n, thusf=n/(T- 1').

equation (III) for K l and f,, for K 3 (third harmonic tuning) and for aflyback ratio of 1/T 0.15. FIG. 3 shows the values of n associated withthe various minimum values. It is found that in order to obtain both Thecurve of FIG. 3 is determined for a flyback EI-IT 1 and sawtooth currentgenerator in which fa satisfies a satisfactory shape of the primaryflyback pulse and a minimum amplitude of the scan oscillations, n ispreferably chosen to be equal to 7 in this case.

It will be evident that the above given proportioning off yields twoimportant advantages.

1. For a given nature and magnitude of the said interfering influencewhich make an exact tuning of the generator impossible, the amplitude ofthe scan oscillations caused thereby is maintained as small as possible.In other words, for a given admitted amplitude of the scan oscillations,considerably larger deviations off and f relative to their exact tuningvalue can be admitted.

2. The spread off caused by tolerances and production spread in thedifferent species of one and the same production series do notsubstantially have any influence on the amplitude of the scanoscillations which is in contrast with the fact when f is located, forexample, in the middle between a maximum and a minimum value, forexample,f (n %/T 1'). Prol duction spreads in f will then have a largeinfluence on the amplitude of the scan oscillations.

It is to be noted that the curve denoted by ER in FIG. 3 is determinedfor a network which in case of a closed switch has a given quantity ofresistive losses in the oscillatory circuit constituted by E, SW, L2, C2(quality factor Q is approximately equal to 20). It is found that whenreducing these losses the maximum values of the ER-curve increase andthe minimum values decrease. By reduction of the resistive losses,particularly by reduction of the copper and iron losses in thetransformer l a further reduction of the scan oscillations can beobtained for the above-mentioned optimum proportioning off In case of anf not proportioned in an optimum manner such a step results in anincrease of the scan oscillation.

in order to obtain an optimum use of the correct proportioning off thequality factor Q of the scan circuit is therefore preferably chosen tobe larger than 25. In case of a correct proportioning off in combinationwith a Q of 25 an oscillation energy ratio (ER) of only approximately /2percent is found for a relatively large (5 percent) deviation of theratio f lf relative to the optimum value of this ratio. This qualityfactor can be simply determined from the expotential decrease of thescan oscillations during the scan period.

tance between a minimum and an adjacent maximum thus is v/(T-rfFurtherrriore FIG. 3 shows that f must deviate not more thanapproximately 4; 1/(T -1') from the optimum n/(T-r) so as to be locatedto a sufficient extent in a minimum value of the ER curve. f YiMLdEQlai/ rm amim h qhoL- ay be approximately usec the admissibledeviation offB is thus approximately :2.3 kHz. It folows from the abovethat forf an absolute accuracy (in kHz) and not a relative accuracy (inpercent) is required.

In the embodiment of FIG. 4 corresponding elements have the samereference numerals as those in FIG. 1. As compared with the embodimentof FIG. 1, FIG. 4 shows an additional transformer winding 5 which isconnected with one side to the positive terminal of the voltage supplysource 7 and with the other side through a parallel LC circuit 16, 17 tothe collector electrode of transistor 12. The equivalent circuit diagramis shown in FIG. 5. in this diagram L3 and C3 mainly represent theinductance of coil 16 and the capacitance of capacitor 17 while L4 isthe leakage inductance between the windings 4 and 5. As alreadyexplained in sa d BE'IQDIEJEEWI appli t n network of 5 has threeresonant frequencies a4, and f,, and a substantially oscillation freescan period can be obtained by choosing these frequencies in such amanner that all three of them satisfy equation (III) for K is an oddinteger, for example K is equal to l, 5 and 7, re-

P9EYX- a, .7 a v In this case the network has two frequencies f,;, and fat which the input impedance of the network is at a minimum (zero) andthe first (f of which is located between fa and f; and the second (f ofwhich is located between fe and f7. Due to the interfering influencesreferred to in the preamble some oscillations will also occur during thescan period in this case. These osciflatio ns will consist of a firstcompo- Rein of the frequency f;;, and a second component of thefrequency f By choosing f within the given limits [i /s. l/(T 'r)] to beequal to an integral multiple of the reciprocal value of the duration ofthe scan period (n/T r with n an integer) the amplitude of the saidfirst component can be adjusted at a minimum and likewise the amplitudeof the said second component can be adjusted at a minimum by choosing fwithin the given limits [iVsH/T- 1-)] to be equal to an integralmultiple of the reciprocal value of the duration of the scan period(m/T- r with m an integer).

It is to be noted that in t l1e equivalent circuit diagram of FIG. 5 anincrease of f;;, in case of an equal location of the other frequenciescan be realized by reducing L3 and L4 and by increasing Cl and C3 and aslight adaptation of the other elements. An increase of f may berealized by reducing C3 and increasing C1 with a slight adaptation ofthe other elements.

What is claimed is:

1. A circuit for generating a sawtooth waveform comprising switchingmeans periodically nonconducting during a flyback period and conductingduring a scan period, a network-means for establishing a plurality ofresonant frequencies having a pair of input terminals coupled to saidswitching means, said network means comprising a transformer having aprimary winding coupled to said switching means and at least onesecondary winding, whereby a leakage inductance exists between saidwindings, said network having a first resonant frequency during saidflyback period substantially equal to the expression wherein K l, S acorrection factor substantially equal to the relative reduction of theslope of the sawtooth waveform at the end of the scan period relative tosaid slope in the middle of said scan period, F said flyback period, T1'= said scan period, said network having a second resonant frequencysubstantially equal to said expression wherein K is an odd integergreater than one, said network having a third resonant frequency betweensaid first and second resonant frequencies at which the impedance atsaid input terminals is a minimum, said third resonant frequency beingsubstantially equal to an integral multiple of the reciprocal of thescan period.

2. A circuit as claimed in claim 1 wherein said net work has a qualityfactor greater than twenty five at said third frequency during said scanperiod.

3. A circuit as claimed in claim 1 wherein said network furthercomprises means for establishing a fourth resonant frequency betweensaid first frequency and said second frequency and being substantiallyequal to said expression for K equal to an odd integer.

4. A circuit as claimed in claim 3 wherein said network furthercomprises means for establishing a fifth resonant frequency between saidfourth and second frequencies at which the impedance at said terminalsis pled between said additional winding and said switch.

UNITED STATES PATENT AND TRADEMARK OFFICE CERTIFICATE OF CORRECTIONPATENT NO. 3, 793 555 DATED February 19, 1974 INVENTOR(S) c li JohannesMaria Van Gils It is certified that error appears in theabove-identified patent and that said Letters Patent are herebycorrected as shown below:

Col. 5', line 20, cancel the formula and insert as follows:

line 35, cancel the formula and insert as follows:

and

err= K7r+ line 41, cancel the formula and insert as follows:

n I 1 T 7 Signed and Scaled this Twenty-eighth Day of September 1976[SEAL] A [res I.

RUTH C. MASON C. MARSHALL DANN Arresting Officer ('ummissium'r nfParenrsand Trademarks UNITED STATES PATENT AND TRADEMARK OFFICE CERTIFICATE OFCORRECTION PATENT NO. 3 793 555 DATED February 19, 1974 INVENTOR(S)Cornelis Johannes Maria Van Gils It is certified that error appears inthe above-identified patent and that said Letters Patent are herebycorrected as shown below:

Col. 5', line 20, cancel the formula and insert as follows:

0t'7'= K7T+ 2 or line 35, cancel the formula and insert as follows:

. '7'i 2 0 (1T- K7T+ W and line 41, cancel the formula and insert asfollows: i

Signed and Scaled this Twenty-eighth Day Of September [SEAL] Anesr:

RUTH C. MASON Arresting Officer

1. A circuit for generating a sawtooth waveform comprising switchingmeans periodically nonconducting during a flyback period and conductingduring a scan period, a network means for establishing a plurality ofresonant frequencies having a pair of input terminals coupled to saidswitching means, said network means comprising a transformer having aprimary winding coupled to said switching means and at least onesecondary winding, whereby a leakage inductance exists between saidwindings, said network having a first resonant frequency during saidflyback period substantially equal to the expression K/2 Tau ( 1 + (4/pi 2K2) . Tau /(T- Tau ) (1 - 2S/3) ), wherein K 1, S a correctionfactor substantially equal to the relative reduction of the slope of thesawtooth waveform at the end of the scan period relative to said slopein the middle of said scan period, Tau said flyback period, T - Tau saidscan period, said network having a second resonant frequencysubstantially equal to said expression wherein K is an odd integergreater than one, said network having a third resonant frequency betweensaid first and second resonant frequencies at which the impedance atsaid input terminals is a minimum, said third resonant frequency beingsubstantially equal to an integral multiple of the reciprocal of thescan period.
 2. A circuit as claimed in claim 1 wherein said network hasa quality factor greater than twenty five at said third frequency duringsaid scan period.
 3. A circuit as claimed in claim 1 wherein saidnetwork further comprises means for establishing a fourth resonantfrequency between said first frequency and said second frequency andbeing substantially equal to said expression for K equal to an oddinteger.
 4. A circuit as claimed in claim 3 wherein said network furthercomprises means for establishing a fifth resonant frequency between saidfourth and second frequencies at which the impedance at said terminalsis a minimum, said fifth frequency being substantially equal to anintegral multiple of the reciprocal of scan period.
 5. A circuit asclaimed in claim 1 further comprising rectifier means coupled to saidsecondary winding for providing a high voltage.
 6. A circuit as claimedin claim 1 further comprising a deflection coil coupled to said primarywinding.
 7. A circuit as claimed in claim 1 wherein said transformerfurther comprises an additional winding, and said circuit furthercomprises a parallel tuned trap coupled between said additional windingand said switch.